1. Field of the Invention
This invention relates to a method for interrogating sensor systems based on wirelessly interrogated passive sensor-transponders as used, for example, for measuring pressure and temperature of air in vehicle tires. More specifically, a preferred embodiment of the invention provides a passive sensor interrogation algorithm which allows high accuracy of measurement of pressure and temperature.
2. The Prior Art
Passive wireless sensors based on resonators have been under development for the last 10 years. They offer a cost-effective batteryless solution for the applications where physical quantities such as temperature, pressure, acceleration, torque etc. need to be measured to be pleasured on rotating or moving parts. If surface acoustic wave (SAW) resonators are employed as sensing elements then their working frequency within the UHF range makes the antenna size (of around 10 cm) suitable for a wide range of practical applications. At the same time a very high Q factor of the SAW resonators around 10000 makes it possible to measure their resonant frequency wirelessly with a good accuracy.
The basic principle behind the SAW resonant sensors is that the resonant frequency depends on the physical quantities mentioned above. Usually the SAW sensing element is designed in such a way that it contains several SAW resonators, each characterised by a different variation of the resonant frequency with temperature, torque. and pressure etc, in other words, different calibration characteristics. The physical quantities are found by measuring wirelessly the resonant frequencies of all the resonators connected to a single sensor antenna and a subsequent solution of simultaneous equations approximating the sensor calibration characteristics.
One application of the passive wireless resonant sensors in the car tire pressure and temperature monitoring system (TPMS) is described in the paper by B. Dixon, V. Kalinin, J. Beckley and R. Lohr “A second generation in-car tire pressure monitoring system based on wireless passive SAW sensors”, Proceedings of 2006 IEEE Frequency Control Symposium. As shown in FIG. 1, the system consists of four SAW sensors 1 (each equipped with its own antenna) installed inside the tires, four interrogation antennas 2 installed under the wheel arches and connected in turn by the RF switch 3 to the input of the interrogation unit 4. The SAW sensing element 1 in this case consists of three SAW resonators with the resonant frequencies occupying a license-free 433 MHz ISM (Industrial, Scientific and Medical) frequency band. By measuring wirelessly the three resonant frequencies one after another the interrogator can determine independently both pressure and temperature inside the tire (one of the resonators is used as a reference to allow differential measurement to minimize influence of parasitic effects such as aging and frequency pulling by a variable antenna impedance).
Wireless measurement of the resonant frequency is adversely affected by two phenomena—noise in the electronic circuits of the interrogator's receiver and RF interference picked up by the interrogation antenna. The noise causes random errors in the measured resonant frequency that may become unacceptably large when the magnitude of the SAW response picked up by the receiver is small (it happens in TPMS at certain angular positions of the wheels). The RF interference may come from another system working in the 433 MHz ISM band such as a remote keyless entry system (RKE) or from the TPMS transmitter of a neighbouring vehicle. It may block the interrogator's receiver and make interrogation impossible. The same problems can also arise in sensing systems for wireless measurement of not only frequency but also phase and group delays (e.g. in delay line sensor systems), modulation depth or other signal parameters. The aim of this application is to disclose an interrogation method that allows improving noise and interference immunity of the wireless interrogator of the passive sensing system.
The resonant frequency can be measured wirelessly by a number of different methods. The method most suitable for the distance of around 1-3 m has been disclosed in the GB patent 2381074 (and corresponding patent U.S. Pat. No. 7,065,459) and GB patent 2411239. The interrogation is performed in the time domain by launching an RF interrogation pulse at the interrogation frequency close to the resonant frequency of the SAW resonator that is being measured, exciting natural oscillation in the resonator, then picking up the natural oscillation after the interrogation pulse is over and analysing its spectrum. The frequency of the natural oscillation corresponding to the maximum of the power spectral density (PSD) is assumed to be equal to the measured resonant frequency.
The interrogator contains a pulsed transmitter (Tx) generating the interrogation pulses at one of the number of possible discrete interrogation frequencies. The pulse width is such that it can efficiently excite the natural oscillations, i.e. it is related to the loaded Q factor of the resonator+the sensor antenna connected to it. In practice it is of the order of 10 μs. The shape of the pulse envelope and its peak power are such that its spectrum complies with the regulations of the country where the sensing system is used. For instance, the envelope can have rise and fall modulated by a Gaussian function to suppress spectrum sidelobes and the peak power should be below 10 dBm in EU countries. The frequency interval between possible interrogation frequencies is also related to the loaded Q factor of the resonator and it is in the range between 25 and 100 kHz at 433 MHz. This arrangement always allows finding such an interrogation frequency that the natural oscillation can be efficiently excited (the spectrum of the interrogation pulse overlaps with the frequency response of the resonator).
The interrogator also contains a wideband superheterodyne receiver (Rx) picking up the natural oscillation after the interrogation pulse is over. The receiver converts the input frequency to an intermediate frequency (IF) that can be easily sampled by an analog-to-digital converter (ADC) but it should be larger than the spectral width occupied by the SAW response. For instance, the nominal IF (the one at the IF output when the input frequency equals the interrogation frequency) can be around 1 MHz. Knowing the frequency of the IF signal, one also knows the frequency of the RF signal at the Rx input. The Rx bandwidth should be wide enough in order to prevent transient processes at the Rx output from corrupting the SAW response. In practice it can be from 1.5 to 5 MHz. The abovementioned documents disclose the use of two IF outputs of the receiver, I(t) and Q(t), shifted relative to each other in phase by 90°. They can be obtained, for instance, by using a quadrature mixer in the Rx frequency down-converter.
The SAW responses I(t) and Q(t) at the IF are sampled by the ADC (samples are taken at the same moments of time both for I and Q channels) during a period of 10 to 20 μs, corresponding to the length of the exponentially decaying SAW response. Then a digital signal processor (DSP) performs a spectral analysis of the signals I(t) and Q(t) and finds precise positions of the maximum of the power spectral density of each signal using parabolic (or any other) interpolation between the calculated spectral lines. The frequencies corresponding to those positions are averaged to give the frequency of the natural oscillation. This method of calculation drastically reduces influence of the unknown initial phase angle of the SAW response on the measured frequency and thus improves the accuracy of the measurement. The alternative approach giving the same high precision is to calculate the power spectral density of the complex signal I(t)+jQ(t) and find the frequency of the natural oscillation as the one corresponding to the maximum of this PSI), again, by using interpolation. It requires fewer calculations in comparison with the previous approach.
The abovementioned documents disclose a general structure of the interrogation algorithm. Interrogation begins with the search phase when the interrogation unit sweeps through all possible interrogation frequencies and finds the ones closest to the resonant frequencies of all three resonators. The documents describe possible ways of finding those optimal interrogation frequencies based on the analysis of the maxima of the calculated PSD values. After the search is complete the measurement phase begins when the resonant frequencies are measured sequentially one after another with high precision. At the measurement phase the interrogation is performed either at the optimum interrogation frequency found during the search or at the interrogation frequency closest to the previously measured resonant frequency (if the measurement is not triggered manually but repeated automatically).
These documents also suggest a number of measures to improve noise and interference immunity of the interrogation unit.
A. Noise
The first measure is to use coherent accumulation of several SAW responses. This is achieved by repeated launching of the interrogation pulses at the same frequency and adding sampled values of I(t) and Q(t) to the values obtained for the previously received SAW response in the DSP buffers. The samples should always be taken at the moments tied to the initial phase angle of the SAW responses to preserve coherency. This can be achieved if the same clock oscillator is used to generate the local oscillator signal in the Rx and the clock frequency in the DSP. Other conditions are disclosed in U.S. Pat. No. 7,065,459. Coherent accumulation of Nc SAW responses improves signal-to-noise ratio by a factor of Nc1/2. Influence of the phase noise of the local oscillator is also reduced by the same factor. In practice, Nc is typically 5 to 30.
The second measure is to average several measurement results for each resonant frequency. Averaging Na frequency readings accumulated in the DSP buffer reduces random errors of the measured frequency by a factor of Na1/2 where, in practice, Na=10 . . . 100. In other words, if σ is the standard deviation of the individual resonant frequency readings fj then the averaged frequency
                              f          ave                =                              1                          N              a                                ⁢                                    ∑                              j                =                1                                            N                a                                      ⁢                          f              j                                                          (        1        )            will have the standard deviationσf ave=σ/√Na.  (2)
An example of the timing diagram for the interrogation pulses radiated by the Tx is shown in FIG. 2. Interrogation begins with the search phase when 19 possible interrogation frequencies are swept. Groups of Nc=6 are launched at each frequency in order to accumulate Nc SAW responses. The distance between the groups is determined by the time needed for the spectral analysis and frequency calculation as well as switching of the Tx to another interrogation frequency. The measurement phase consists of three groups of pulses at the optimal interrogation frequencies fi1, fi2, fi3 repeated Na times in order to fill in the three buffers of the measured resonant frequencies f1,2,3. The average frequencies can be calculated at the end of those 3Na groups if the measurement needs to be performed once, or the cycle can be repeated if the average frequency readings need to be periodically updated. Average frequencies can also be calculated as moving averages.
The problem with this approach is that the amplitudes of the SAW responses are usually not constant; for example, in the case of TPMS, they vary as a result of rotation of the wheels and thus variation of the distance between the interrogation antenna and the sensor antenna as well as variation of the sensor antenna impedance. For certain positions the amplitude of the SAW response is so small that the frequency reading obtained by the reader becomes unreliable. According to the method proposed in the abovementioned documents these readings need to be disregarded, i.e. excluded from calculation of the average frequency in Eq. (1). As a result, the number of the groups of interrogation pulses needed to all in the averaging buffers can be larger than 3Na by perhaps 25% to 50%. This increases the minimum achievable data update period. In practical TPMS, filling in the buffers can take up to 300 . . . 500 ms for Na=40. As a result, temperature and pressure cannot be updated faster than 1.2 . . . 2 s in the case of the measurement performed in four wheels. For some applications, in particular in motor sport, this update period is too large. One aim of the invention is to provide a method that can either reduce random errors of the wireless resonant frequency measurement for a fixed update period or reduce the update period for a given standard deviation of the random errors.
B. Interference
Broadband interference has the same effect on the measurement as noise. Narrowband interference is more dangerous because it may completely spoil the measurement results or cause an unacceptably large systematic error if its frequency is within approximately ±100 kHz around the measured resonant frequency and the amplitude is above a certain threshold level (for instance 20 dB below a typical SAW response amplitude). The abovementioned documents adopt the strategy similar to the one used in CSMA-CD (Carrier-Sense Multiple Assess with Collision Detection) communication protocols. Before launching the interrogation pulse at the beginning of coherent accumulations, the interrogator's receiver “listens” to the interference by taking samples of the IF signal. If interference is detected, the measurement is delayed by a random interval of time. Since the interference needs to be detected only within a limited frequency range of about 200 kHz, much smaller than the Rx bandwidth, detection of the narrowband interference cannot be performed by a simple measurement of the rms or peak value of the IF signal as it is usually done in communication systems. Instead, it is proposed to calculate PSD values of the IF signal within the frequency range of interest and compare the maximum PSD value with the threshold. Detection of interference within the limited bandwidth allows increasing capacity of the system by a factor of three (in the case where there are three SAW resonators in the sensor) because it makes possible simultaneous interrogation of three resonators by three different systems.
The abovementioned documents do not disclose how long the Rx should “listen” to the interference. If a source of interference uses the ISM band for a slow digital data transmission by means of frequency-shift keying (FSK) signals or analogue transmission by means of AM or FM signals then the period of “listening” is not critical, it can be quite short—the interference will not be missed. A more dangerous situation is when the interference is generated by a neighbouring TPMS system of the same type installed on another vehicle. In this case the “listening” period Tl should be as long as the period Ti of the interrogation pulses which is approximately from 20 to 40 μs. It is determined by the length of the interrogation pulse 5 and the length of the SAW response sampling window 6 as shown in FIG. 3. Only in this case will the interference sampling window 7 intercept the whole single interrogation pulse and correctly determine its maximum PSD.
The problem with this approach is that the number of interference samples in this case will exceed the number of the signal samples (the sampling period in both windows should be the same to be able to use the same sine and cosine look-up table and the same routine for calculation of the PSD in the DSP). As a result, the size of the sine and cosine look up table stored in the DSP memory has to be significantly increased, which may pose a problem for inexpensive DSP chips used in the SAW interrogation units. Another aim of the invention is to provide a method of improving robustness of the interference detection and reducing the required number of interference samples.